Synchronous motor driving system and sensorless control method for a synchronous motor

ABSTRACT

Axial error calculation unit is provided for estimating an axial error Δθ between a d-q axis and a dc-qc axis by using Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of all rotational speeds except zero of a rotational speed command of a synchronous motor, Ld denoting an inductance on a magnetic pole axis d of the synchronous motor, Lq an inductance on a q axis orthogonal to the magnetic pole axis d, Ke a generated power constant of the motor, Id* a current command of the d axis, Iq* a current command on a q axis, Idc a detected current value on an assumed dc axis on control, and Iqc a detected current value on an assumed qc axis orthogonal to the assumed dc axis. Irrespective of presence of saliency, position sensorless control can be achieved in a wide range a low to high speed zone.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates to a synchronous motor drivingsystem, and more particularly, it relates to a control method forachieving a highly precise and high-performance synchronous motordriving system without using any sensors for detecting a rotationalspeed of a synchronous motor and a position of a magnetic pole.

[0003] 2. Description of the Related Art

[0004] Many developments have been made for a method for controlling asynchronous motor without detecting the rotational speed of thesynchronous motor and the position of a magnetic pole. Such controlmethods are usually classified into two types.

[0005] A first type is a control method based on speed/position sensorvector control of the synchronous motor. Instead of using thespeed/position sensor, a magnetic pole position estimating instrument,and a speed estimating instrument are used. For example, a method isknown, which is described in a document 1: “No. SPC-00-67: A NewPosition Sensorless Control of IPM Synchronous Motor using DirectPosition Error Estimation”, by inventors Kiyoshi Sakamoto, YoshitakaIwazi, and Tsunehiro Endo, in “IEEJ Semiconductor PowerConversion/Industrial Electric Power Application Joint ResearchMaterial” (11, 2000). This method is known as a vector controlsensorless system.

[0006] A second type is a control method called a V/F control system,which controls the synchronous motor by an open loop.

[0007] In the case of the vector control sensorless system, except fornon-presence of a position/speed sensor, a configuration itself of thecontrol system is similar to that of a vector control system equippedwith a sensor. Accordingly, a high-performance synchronous motor drivingsystem can be achieved.

[0008]FIG. 15 shows a relation between a d-q coordinate axis and anassumed axis dc-qc by a vector with a magnetic pole axis of asynchronous motor set as a reference axis. For vector control, as shownin FIG. 15, a magnetic pole axis of the synchronous motor is set as a daxis, an axis orthogonal to the same as a q axis. Then, by properlycontrolling a voltage and a current applied to the synchronous motor oneach axis, high-performance making utmost use of synchronous motorperformance is achieved. According to this vector control sensorlesssystem, torque can be made linear, and efficiency can be maximized.

[0009] In the case of the vector control sensorless system, a dc-qc axisis set by assuming a d-q axis on control, deviation (axial error) Δθfrom a real d-q axis is estimated, and a dc-qc axial phase is adjustedto reduce the deviation to zero. Thus, in the case of the vector controlsensorless system, a method of estimating an axial error Δθ is a mostimportant factor for deciding control performance.

[0010] In well-known examples, several estimation methods of axialerrors AO have been presented according to the rotational speed zones ofthe synchronous motor. In practice, all the speed zones are covered byusing these control methods in association.

[0011] On the other hand, in the case of the V/F control system, nospeed or current automatic adjustment units are provided, and a voltageto be applied to the synchronous motor is decided. As its conventionalexample, a control method is described in JP-A-2000-236694. In the caseof the V/F control, different form the case of the vector sensorlesssystem, a magnetic pole axis is not estimated. Thus, a configuration ofa control system is greatly simplified. However, if a load is suddenlychanged during driving, transient vibration may occur. In order tosuppress such transient vibration, JP-A-2000-236694 presents a controlsystem for correcting a speed based on a current detected value.

[0012] In the case of the vector control sensorless system, a sensorlesssystem must be switched to another according to a driving speed of thesynchronous motor. The method described in the document 1 estimates anaxial error Δθ based on a speed electromotive voltage of the synchronousmotor in principle, which can be achieved only in a middle/high speedzone. A similar problem is inherent in the method described inJP-A-8-308286.

[0013] On the other hand, as a sensorless system of a low speed, manymethods have been presented, which uses a inductance difference(saliency) of the synchronous motor. For example, as described inJP-A-7-245981, there is a method for superposes a higher harmonic waveon a voltage command, and calculates an axial error based on a higherharmonic current component thus generated.

[0014] In this method, however, since it is necessary to superpose thehigher harmonic wave, a pulsation component is generated in asynchronous motor current, causing a considerable reduction inefficiency of the synchronous motor. In addition, because of thesuperposed wave, electromagnetic noise is increased. Especially, todetect an axial error with high sensitivity, the amount of superposedwaves to be injected must be increased, and thus it is difficult tosolve the above-described problems, and achieve high control performanceat the same time.

[0015] In addition, since the saliency of the synchronous motor is used,the system cannot be applied to a synchronous motor of a non-saliencytype. Further, when the low-speed system, and the middle/high speedsystem are used in association, the systems must be switched accordingto a speed, and thus shocks occur following the switching.

[0016] On the other hand, in the case of the V/F control, thesynchronous motor can be driven from a low to high speed zone by aconfiguration of a single control system.

[0017] However, in the V/F control, since the d-q axis in thesynchronous motor is not basically coincident with the dc-qc axis oncontrol, it is difficult to achieve high-performance control. Forexample, it is difficult to achieve high-speed response to a change in arotational speed command, linear control of torque, maximum efficiencycontrol and the like. Accordingly, there is a possibility that externaldisturbances such as fluctuation in load torque may cause inconveniencessuch as vibration or excessive current.

SUMMARY OF THE INVENTION

[0018] An object of the present invention is to provide a synchronousmotor driving system equipped with means for stabilizing a controlsystem without superposing any higher harmonic waves, controlling alow-speed zone to a high-speed zone by a continuous method, andachieving a vector control sensorless system.

[0019] The foregoing object of the present invention can be achieved byproviding means for calculating an error Δθ as deviation between amagnetic pole axis in a synchronous motor and a magnetic pole axis oncontrol, as a function of a driving current command, a detected currentvalue, an inductance constant, and generated power constant of thesynchronous motor, calculating an axial error by applying the axialerror calculating means to all speed zones of the synchronous motorexcept zero, and correcting the magnetic pole axis on the control basedon the axial error.

[0020] That is, in order to achieve object, in accordance with thepresent invention, there is provided a synchronous motor driving systemwhich comprises a synchronous motor, an inverter for driving thesynchronous motor, a rotational speed command generator for supplying arotational speed command to the synchronous motor, and a control unitfor calculating a voltage applied to the synchronous motor, saidsynchronous motor driving system comprising axial error calculationmeans for estimating an axial error Δθ between a d-q axis and a dc-qcaxis by using Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of allrotational speeds except zero of the rotational speed command of thesynchronous motor wherein Ld is an inductance on a magnetic pole axis d,Lq is an inductance on a q axis orthogonal to the magnetic pole axis d,Ke is a generated power constant of the motor, Id* is a current commandof the d axis, Iq* is a current command on the q axis, Idc is a detectedcurrent value on an assumed dc axis on control, and Iqc is a detectedcurrent value on an assumed qc axis orthogonal to the assumed dc axis;and means for adjusting the dc-qc axis to the d-q axis based on thecalculated value of the axial error Δθ.

[0021] Furthermore, in order to achieve the above object, in accordancewith the present invention, there is provided a synchronous motordriving system which comprises a synchronous motor, an inverter fordriving the synchronous motor, a rotational speed command generator forsupplying a rotational speed command to the synchronous motor, and acontrol unit for calculating a voltage applied to the synchronous motor,said synchronous motor driving system comprising axial error calculationmeans for estimating an axial error Δθ as a function of an inductance Land a generated power constant Ke among the resistance R, the inductanceL and a generated power constant Ke as synchronous motor constants ofthe synchronous motor in a range of all rotational speeds except zero ofa rotational speed command of the synchronous motor; and means foradjusting the dc-qc axis to the d-q axis based on the calculated valueof the axial error Δθ, wherein Ld is an inductance on a magnetic poleaxis d, Lq is an inductance on a q axis orthogonal to the magnetic poleaxis d, Ke is a generated power constant of the motor, Id* is a currentcommand of the d axis, Iq* is a current command on the q axis Iq*, Idcis a detected current value on an assumed dc axis on control, and Iqc isa detected current value on an assumed qc axis orthogonal to the assumeddc axis.

[0022] The axial error calculation means can be adapted to calculate anaxial error Δθ by using current commands Id* and Iq* on the d-q axis,and detected current values Idc and Iqc on the dc-qc axis, wherein Ld isan inductance on a magnetic pole axis d, Lq is an inductance on a q axisorthogonal to the magnetic pole axis d, Ke is a generated power constantof the motor, Id* is a current command of the d axis, Iq* is a currentcommand on the q axis, Idc is a detected current value on an assumed dcaxis on control, and Iqc is a detected current value on an assumed qcaxis orthogonal to the assumed dc axis.

[0023] In this case, instead of the detected current value Idc on the dcaxis, a current command Id* can be used.

[0024] In addition, the synchronous motor driving system may be providedwith means for detecting a DC current on a power source side of theinverter, and synchronous motor current estimating means for estimatingan AC current of the synchronous motor based on the detected DC currentand a driving pulse signal for driving the inverter, and the axial errorΔθ may be calculated using the estimated current as a detected currentvalue.

[0025] The synchronous motor driving system can be provided with meansfor detecting a DC current on a power source side of the inverter, andIqc estimating means for estimating a current value on the qc axis ofthe synchronous motor based on the detected DC current, and a detectedvalue or a set value of a DC voltage of the inverter, and the axialerror Δθ can be calculated using the estimated current as a detectedcurrent value.

[0026] In any of the above-described synchronous motor driving systems,in the calculation of the axial error Δθ, a correction term may beprovided to make correction in accordance with a rotational speedcommand of the synchronous motor, and the correction term may be set asa function of weight which increases as the rotational speed commandapproaches zero.

[0027] The current command Iq* on the q axis can be made based on thedetected current value or the estimated value on the qc axis.

[0028] The synchronous motor driving system can be provided with meansfor estimating a speed deviation between the rotational speed commandand a real rotational speed based on the calculated value of the axialerror Δθ, and the q axis current command Iq* of the synchronous motorcan be made based on the estimated value of the speed deviation.

[0029] The synchronous motor driving system may be provided with meansfor estimating a rotational speed of the synchronous motor based on thecalculated value of the axial error Δθ, and the q axis current commandIq* of the synchronous motor may be made based on a deviation betweenthe estimated value and the rotational speed command.

[0030] In the above-described synchronous motor driving systems of thethree types, the current command Id* of the d axis can be made based onthe current command Iq* of the q axis.

[0031] In the present invention, the synchronous motor may be a salienttype or a non-salient type.

[0032] Other objects, features and advantages of the invention willbecome apparent from the following description of the embodiments of theinvention taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0033]FIG. 1 is a block diagram showing a configuration of a synchronousmotor driving system according to a first embodiment of the presentinvention.

[0034]FIG. 2 is a perspective view showing a schematic structure of thesynchronous motor driving system of the first embodiment.

[0035]FIG. 3 is a block diagram showing an internal configuration of avoltage command calculator 12 in the synchronous motor driving system ofthe first embodiment of the present invention.

[0036]FIG. 4 is a block diagram showing an internal configuration of anaxial error calculator 14 in the synchronous motor driving system of thefirst embodiment of the present invention.

[0037]FIG. 5 is a block diagram showing a configuration of a synchronousmotor driving system according to a second embodiment of the presentinvention.

[0038]FIG. 6 is explanatory view showing an operation of a synchronousmotor current estimator 18 in the synchronous motor driving system ofthe second embodiment of the present invention.

[0039]FIG. 7 is a block diagram showing a configuration of a synchronousmotor driving system according to a third embodiment of the presentinvention.

[0040]FIG. 8 is a block diagram showing an internal configuration of anaxial error calculator 14D in a synchronous motor driving systemaccording to a fourth embodiment of the present invention.

[0041]FIG. 9 is an explanatory view showing an operation of a functiongenerator in the synchronous motor driving system of the fourthembodiment of the present invention.

[0042]FIG. 10 is a block diagram showing an internal configuration of acontrol unit 2E in a synchronous motor driving system according to afifth embodiment of the present invention.

[0043]FIG. 11 is a block diagram showing an internal configuration of acontrol unit 2F when the synchronous motor driving system of the fifthembodiment of the present invention is applied to the third embodimentshown in FIG. 7.

[0044]FIG. 12 is a block diagram showing an internal configuration of acontrol unit 2G in a synchronous motor driving system according to asixth embodiment of the present invention.

[0045]FIG. 13 is a block diagram showing an internal configuration of acontrol unit 2H in a synchronous motor driving system according to aseventh embodiment of the present invention.

[0046]FIG. 14 is a block diagrams showing an internal configuration of acontrol unit 2J in a synchronous motor driving system according to aneight embodiment of the present invention.

[0047]FIG. 15 is a view showing a relation between a d-q coordinate axiswith a magnetic pope axis of a synchronous motor set as a reference, andan assumed dc-qc axis on control by a vector.

DESCRIPTION OF THE EMBODIMENTS

[0048] Next, description will be made of the preferred embodiment of thesynchronous motor driving system of the present invention with referenceto FIGS. 1 to 15.

[0049] First Embodiment

[0050]FIG. 1 is a block diagram showing a configuration of a synchronousmotor driving system according to a first embodiment of the presentinvention. The synchronous motor driving system of the first embodimentcomprises the following elements: a rotational speed command generator 1for supplying a rotational speed command ωr* to the synchronous motor; acontrol unit 2 for calculating amplitude, a frequency and a phase of avoltage applied to the synchronous motor; a pulse width modulation (PWM)generation 3 for generating a pulse for driving an inverter 4 based on avoltage command V1*; the inverter 4 for driving the synchronous motor:the synchronous motor 5 to be controlled; a current detector 6 fordetecting a current of the synchronous motor 5; a conversion gain 7 forconverting the rotational speed command ωr* into an electrical anglefrequency command ω1* of the synchronous motor with P set as a pole; anintegrator 8 for calculating an AC phase θc in the control unit; a dqcoordinate converter 9 for converting a current value on a three-phaseAC axis into a component on a dc-qc axis as a rotation coordinate axis;an Iq* generator 10 for supplying a current command Id* of a d axiscomponent of the synchronous motor; an Iq* generator 11 for supplying acurrent command Iq* of a q axis component (torque component) of thesynchronous motor; a voltage command calculator 12 for calculatingvoltage commands Vdc* and Vqc* on a dc-qc axis based on ω1*, Id* andIq*; a dq reverse converter 13 for converting the voltage commands Vdc*and Vqc* on the dc-qc axis into values on the three-phase AC axis; anaxial error calculator 14 for estimating an axial error between a d-qaxis and a control axis dc-qc of the synchronous motor; a zero generator15 for supplying a zero command to the axial error; an adder 16 foradding or subtracting a signal; and a magnetic pole axis estimation gain17 for calculating an amount of correction to the electrical anglefrequency command ω1* by using the axial error. The inverter 4 includesa DC power source unit 41 constituting a main circuit power source ofthe inverter 4, a main circuit unit 42 of the inverter, a gate driver 43for generating a gate signal to the main circuit; a three-phase AC powersource 411 for supplying power to the inverter 4, a diode bridge 412 forrectifying the three-phase AC power source, and a smoothing capacitor413 for suppressing a pulsation component contained in a DC powersupply.

[0051]FIG. 2 is a perspective view showing a schematic structure of thesynchronous motor driving system of the first embodiment shown inFIG. 1. The synchronous motor driving system of the present inventionmainly has an AC power source unit, a control/inverter unit, and asynchronous motor. As shown in FIG. 2, a control board includes therotational speed command generator 1, the control unit 2, and the PWMgenerator 3, which are all shown in FIG. 1. Actually, the control boardis a digital circuit around a micro-processor. Also, the inverter maincircuit 4, the current detector 6 and the like are installed in oneunit.

[0052] Next, description is made of an operation principle of the firstembodiment by referring to FIG. 1. The conversion gain 7 calculates anelectrical angle frequency ω1* of the synchronous motor based on arotational speed command ωr* from the rotational speed command generator1, and outputs it.

[0053] On the other hand, in the axial error calculator 14, an axialerror Δθ is estimated based on a current command and a detected currentvalue. The magnetic pole axis estimation gain 17 calculates a speedcorrection amount Δω1 based on the estimated axial error Δθ. The adder16 adds the ω1* and the Δω1 together to obtain ω1c. The phase calculator8 integrates the ω1c to obtain an AC phase θc in the control unit. As aresult, the AC phase θc is corrected by the Δθc. Then, bycoordinate-converting the detected value of the three-phase AC currentbased on this θc, an Idc as a dc axis component, and an Iqc as a qc axiscomponent are obtained. At the Id* generator 10 and the Iq* generator11, current commands of respective axis components of the synchronousmotor 5 are supplied. A method of generating the Id* and the Iq* will bedescribed in detail later.

[0054] The voltage command calculator 12 calculates voltages Vdc* andVqc* to be applied to the synchronous motor 5 based on the rotationalspeed ω1* and the current commands Id* and Iq* by an equation (2). Inthe equation, R: motor resistance, Ld: d axis inductance, Lq: q axisinductance, and Ke: generated power constant of motor. [Equation 2]

V _(dc) *=R·I _(d) *−ω ₁ L _(q) ·I _(q)*

V _(qc) *=ω ₁ ·L _(d) ·I _(d) *+R·I _(q) *+I _(e)·ω₁

[0055] The equation (2) is similar to a calculation equation used fornormal vector control. The equation (2) is described, for example as anequation (4.6) in a document 2: p 78, “Theory and Designing Practice ofAC Servo System”, by Hidehiko Sugimoto, Sogo Denshi Publishing (May,1990).

[0056] The dq reverse converter 13 coordinate-converts the voltages Vdc*and Vqc* obtained by the equation (2) into voltage command values V1* onthe three-phase AC axis. Then, at the PWM generator 3, a voltage commandV1* is converted into a pulse width. The gate driver 43 drives aswitching element based on this pulse signal, and applies a voltageequal to each of the voltages Vdc* and Vqc* to the synchronous motor 5.

[0057]FIG. 3 is a block diagram showing an internal configuration of thevoltage command calculator 12 in the synchronous motor driving system ofthe first embodiment of the present invention. The voltage commandcalculator 12 includes a gain 121 equivalent to a resistance value (R)of the synchronous motor, a gain 122 equivalent to a d axis inductance(Ld), a gain 123 equivalent to a q axis inductance (Lq), a multiplier124, and a gain 125 equivalent to a generated power constant (Ke).

[0058] As shown in the equation (2) and FIG. 3, a voltage command iscalculated by using constants R, Ld, Lq and Ke of the synchronous motor.If these constants of the synchronous motor are accurate, then thesynchronous motor is driven at a rotational speed indicated by a commandvalue, and a current value.

[0059] By the axial error calculator 14, and the magnetic pole axisestimation gain 17, a phase locked loop (PLL) is formed and, bycorrecting the ω1*, a phase angle θc is corrected. As a result, an axialerror Δθ is controlled to zero. Control response time for converging theaxial error to zero is decided by set response by the magnetic pole axisestimation gain 17. In addition, the magnetic pole axis estimation gainmay be a proportional gain in the case of the configuration of thecontrol system of FIG. 1

[0060] Next, description is made of an operation of the axial errorcalculator 14 as a feature of the present invention. According to thedocument 1, an axial error Δθ can be calculated by an equation (3) usinga voltage command, a detected current value, and a constant of thesynchronous motor. $\begin{matrix}{{\Delta\theta} = {\tan^{- 1}\frac{V_{d\quad c}^{*} - {R \cdot I_{d\quad c}} + {\omega_{1}{L_{q} \cdot I_{q\quad c}}}}{V_{q\quad c}^{*} - {R \cdot I_{q\quad c}} - {\omega_{1}{L_{q} \cdot I_{d\quad c}}}}}} & \lbrack {{Equation}\quad 3} \rbrack\end{matrix}$

[0061] Codes Vdc* and Vqc* in the equation (3) represent parametersshown in the equation (2), both of which strongly depend on anelectrical angle frequency ω1. If Id* and Iq* are constant, then Vdc*and Vqc* are changed substantially in proportion to ω1*. Thus, when theω1* is near zero, a denominator/numerator of the equation (3) approacheszero, causing a considerable reduction in calculation accuracy.

[0062] When the ω1* is near zero, dependence of a term of resistance Ris increased. Since the resistance R is strongly affected by temperaturedependence, nonlinearity by a semiconductor device, or the like,accurate setting of the resistance R is difficult, making it impossibleto establish the equation (3). Consequently, it is very difficult toestimate an axial error in a wide speed range by using the equation (3).An equation (4) is obtained by substituting the equation (2) for theequation (3), and arranging it. $\begin{matrix}{{\Delta\theta} = {\tan^{- 1}\frac{{\omega_{1}{L_{q}( {I_{q\quad c} - I_{q}^{*}} )}} - {R( {I_{d\quad c} - I_{d}^{*}} )}}{{K_{e} \cdot \omega_{1}} - {\omega_{1}( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} )} - {R( {I_{q\quad c} - I_{q}^{*}} )}}}} & \lbrack {{Equation}\quad 4} \rbrack\end{matrix}$

[0063] In a denominator/numerator of the equation (4), it can be seenthat there are two types of terms, i.e., for ω1 and R.

[0064] An R term of the numerator (=−R(Idc-Id*)) indicates an amount ofvoltage reduced according to deviation between Idc and Id*. The Id isnormally controlled to zero in the synchronous motor of a non-salienttype. Also, in the case of that of a salient type, in a stationarystate, changes only occur in a range of 20 to 30% of synchronous motorrating. Thus, the amount of reduced voltage of this term becomes a smallvalue of about 1% or lower, which can be ignored.

[0065] An R term of the denominator (=−R(Iqc-Iq*) indicates a voltagereduction of 1% or lower even in rated current, which can be ignoredsubstantially in all speed range compared with a term of Ke·ω1.Moreover, as higher efficiency is strongly demanded for the synchronousmotor in recent years, the resistance R of the synchronous motor tendsto be designed smaller and smaller.

[0066] For the above-described reasons, by ignoring the R term in theequation (4), the equation (4) can be simplified to be an equation (5).That is, assuming that a d axis inductance is Ld[H], a q axis inductanceLq[H], a generated power constant of the motor Ke[Wb], a current commandvalue of a magnetic pole d axis Id*, a current command on a q axisorthogonal to the magnetic pole d axis Iq*, a detected current value onthe assumed dc axis of the magnetic pole axis Idc, and a detectedcurrent value on the qc axis orthogonal to the dc axis Iqc, an axialerror Δθ is calculated by the equation (5) using the current commandsId* and Iq* on the d-q axis, and the detected current values Idc, Iqc onthe dc-qc axis. $\begin{matrix}{{\Delta\theta} = {\tan^{- 1}\frac{L_{q}( {I_{q\quad c} - I_{q}^{*}} )}{K_{e} - ( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} )}}} & \lbrack {{Equation}\quad 5} \rbrack\end{matrix}$

[0067] In the equation (5), the ω1 term is cancelled, no electricalangle frequencies are present, and an axial error Δθ can be calculated.As the resistance of the equation (4) is ignored, an estimation error isgenerated in a very low speed zone of 1 to 2% or lower. However, it canbe detected whether there is an axial error. Even if there is a smallestimation error, since an axial error Δθ can be reduced to zero at theend, a vector control sensorless system can be achieved in a range ofsubstantially all speeds.

[0068] In the very low speed zone of 1 to 2% or lower, estimation of anaxial error is inevitably difficult. Thus, for example, it is impossibleto output rated torque at a zero speed. However, passage can be allowedthrough the very low speed zone during acceleration/deceleration of thesynchronous motor.

[0069]FIG. 4 is a block diagram showing an internal configuration of theaxial error calculator 14 in the synchronous motor driving system of thefirst embodiment of the present invention. That is, FIG. 4 shows theconfiguration of the axial error calculator 14 using the equation (5).The axial error calculator 14 includes a generated power constant setter126, and an arc tangent calculator 127.

[0070] The generated power constant setter 126 outputs a generated powerconstant (Ke), and the arc tangent calculator 127 calculates an arctangent of Y0/X0 for two inputs X0 and Y0. As shown in FIG. 4, byentering current commands Id* and Iq*, and detected current values Idcand Iqc, it is possible to calculate an axial error Δθ without anydependence on ω1.

[0071] In the equation (5), by using an approximation of tan(x)=x in arange of small x, an axial error can be calculated in approximation byan equation (6). $\begin{matrix}{{\Delta\theta} = \frac{L_{q}( {I_{q\quad c} - I_{q}^{*}} )}{K_{e} - ( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} )}} & \lbrack {{Equation}\quad 6} \rbrack\end{matrix}$

[0072] In addition, in the case of the synchronous motor of thenon-salient type, in the equation (5) or (6), by replacing Ld and Lq byone inductance, an axial error can be calculated more easily. In thecase of the synchronous motor of the non-salient type, since Id isnormally controlled to be zero, calculation can be simplified more bysetting Id* to Id*=0 in each calculation equation.

[0073] Next, more detailed description is made of the equations (5) and(6) to simplify the axial error calculation equation more. Terms ofnumerators in both equations relate to a torque current Iq, and aregreatly changed by load fluctuation or the like of the synchronousmotor. In other words, it can be understood that the numerator termsgreatly affect presence of axial errors and error directions(polarities).

[0074] On the other hand, terms of denominators relate to a generatedpower constant Ke and Id. A code Ke denotes a magnetic flux of apermanent magnet, which is larger than that generated by Id. Thus, Ke isdominant normally. Therefore, even if Id is slightly changed duringtransition, great fluctuation of a denominator value may be limited.Therefore, when priority is placed on simplicity of calculation ratherthan on precise axial error calculation, Id* can be used instead of Idc.In such a case, the equation (5) is changed to an equation (7).$\begin{matrix}{{\Delta\theta} = {\tan^{- 1}\frac{L_{q}( {I_{q\quad c} - I_{q}^{*}} )}{K_{e} - {( {L_{q} - L_{d}} )I_{d}^{*}}}}} & \lbrack {{Equation}\quad 7} \rbrack\end{matrix}$

[0075] By using the equation (7), an axial error can be calculated moreeasily, making it possible to reduce a load on control calculation.

[0076] Second Embodiment

[0077]FIG. 5 is a block diagram showing a configuration of a synchronousmotor driving system according to a second embodiment of the presentinvention. The synchronous motor driving system of the second embodimentincludes an inverter 4B, a DC current detector 44 for detecting acurrent I0 flowing from a DC power supply unit 41 to an inverter maincircuit unit, and a synchronous motor current estimator 18 forestimating a synchronous motor current based on a detected value of thecurrent I0.

[0078] In the second embodiment of FIG. 5, the current detector 6 isremoved from the configuration of the system shown in FIG. 1, andinstead the DC current detector 44, and the synchronous motor currentestimator 18 are added. The DC current detector 44 detects a current byusing HALL CT or shunt resistance. The synchronous motor currentestimator 18 estimates a synchronous motor current based on the detectedvalue of the current IO, and a pulse waveform outputted from a PWMgenerator 3.

[0079] A control unit 2 controls the synchronous motor by regarding anestimated value Ilc as a detected current value of the synchronousmotor. Thus, an operation of the control unit 2 itself is similar tothat of the first embodiment.

[0080]FIG. 6 is explanatory view showing an operation of the synchronousmotor current estimator 18 in the synchronous motor driving system ofthe second embodiment of the present invention. In FIG. 6, each of (a)to (c) shows a PWM pulse waveform of each phase. At 1, a switch (Sup,Svp or Swp) of a plus side is turned ON and, at 0, a switch (Sun, Svn orSwn) of a minus side is turned ON.

[0081] Now, assuming that a synchronous motor current is similar to thatshown in FIG. 6(d), the DC current IO of the inverter has a waveformsimilar to that shown in FIG. 6(e). For the waveform of FIG. 6(e), thereare four modes described below.

[0082] (1) Mode 1:

[0083] Sup=ON, Svp=ON, Swp=ON→I0=0

[0084] (2) Mode 2:

[0085] Sup=ON, Svp=ON, Swp=OFF→I0=Iu+Iv=−Iw

[0086] (3) Mode 3:

[0087] Sup=ON, Svp=OFF, Swp=OFF→I0=Iu

[0088] (4) Mode 4:

[0089] Sup=OFF, Svp=OFF, Swp=OFF→I0=0

[0090] Thus, Iu can be detected if the DC current I0 is detected in theswitching state of the mode 3 and, in the state of the mode 2, Iw can bedetected. Iv may be calculated based on Iu and Iw. A basic operation ofthe synchronous motor current estimator is similar to that of a methoddisclosed in, for example JP-A-6-153526 or JP-A-8-19263.

[0091] However, there was a big problem when this synchronous motorcurrent estimator was used for the conventional vector controlsensorless system. As described above, in order to achieve the vectorcontrol sensorless system in a low-speed zone, the method using thesaliency of the synchronous motor, i.e., the method of estimating anaxial error from a higher harmonic wave component of a current bysuperposing a higher harmonic wave on a voltage, is only available.Thus, it was necessary to accurately detect a higher harmonic waveflowing to the synchronous motor.

[0092] In the configuration of the control system of FIG. 5, as shown inFIG. 6, because of dependence of a timing for current detection on astate of the PWM pulse, it is difficult to accurately detect a higherharmonic wave component in a current. A basic wave component can bedetected without any problems, since a frequency is sufficiently lowercompared with a PWM pulse frequency. Accordingly, detection of thehigher harmonic wave is a problem. To increase accuracy of detection,improvements such as an increase in the amount of superposing a higherharmonic wave, or a reduction in a frequency of the superposed wave.Either case may reduce efficiency, causing a great increase in noise.

[0093] On the other hand, in the axial error calculation of the presentinvention, it is not necessary to superpose any higher harmonic wavesand, by using the equation (5) or (6), accurate axial error calculationis possible in substantially all speed zones. Therefore, ahigh-performance can be expanded to substantially all the zones.

[0094] Third Embodiment

[0095] The above-described second embodiment enables the number ofcurrent sensors to be reduced, and thus provides an advantage ofsimplifying the configuration of the control system. However, thefollowing problems are inherent. That is, when a rotational speed of thesynchronous motor is low, and an output voltage is small, the periods ofthe modes 2 and 3 shown in FIG. 6 become short, necessitating reading ofa very narrow pulse-like current. Waveforms in FIG. 6 are for principleexplanation, and I0 represents a staircase having no vibration. Inpractice, however, ringing is superposed in a current waveform followingswitching. If a pulse width is narrow, this effect cannot be ignored.

[0096]FIG. 7 is a block diagram showing a configuration of a synchronousmotor driving system according to a third embodiment of the presentinvention, which is provided to solve the problems of the secondembodiment. The synchronous motor driving system of the third embodimentincludes a control unit 2C, a filter 19 for removing a pulsationcomponent contained in a DC current I0, and an Iqc estimator 20 forestimating a q axis current Iqc on a control axis.

[0097] The third embodiment of FIG. 7 is substantially similar in systemconfiguration to the second embodiment of FIG. 5. However, thesynchronous motor current estimator 18 of FIG. 5 is removed and,instead, the filter 19 and the Iqc estimator 20 are added.

[0098] In the control unit 2C, compared with the first embodiment ofFIG. 1, the dq coordinate converter 9 is removed, and a systemconfiguration is employed, where Idc and Iqc are not calculated from asynchronous motor current.

[0099] Instead, Iqc necessary for control is obtained by using the Iqcestimator 20. No detection/estimation of Idc is carried out.Accordingly, an axial error calculator 14 estimate an axial error Δθaccording to the equation (7) without using Idc.

[0100] Next, description is made of an operation principle for thefilter 19 and the Iqc estimator 20. The filter 19 removes a PWM pulsecomponent from a DC current I0, and extracts an average value of I0.This filter is provided for the purpose of removing a carrier frequencycomponent. Accordingly, a cutoff frequency of the filter only needs tobe set equal to about 1 of several, or 1 of several tens of a carrierfrequency. Thus, an effect of ringing following switching can becompletely removed. As a result, from an output of the filter 19, a DCcurrent I0 having a higher harmonic wave removed is obtained. The Iqcestimator 20 estimates Iqc by using the current IO having the higherharmonic wave removed.

[0101] Next, description is made of a principle of the Iqc estimation. Arelation between a voltage/current on a d-q axis of the synchronousmotor, and a DC power supply voltage VO and the DC current IO of aninverter is represented by an equation (8) with regard to power.$\begin{matrix}{{I_{0}V_{0}} = {\frac{3}{2}( {{V_{d}I_{d}} + {V_{q}I_{q}}} )}} & \lbrack {{Equation}\quad 8} \rbrack\end{matrix}$

[0102] A coefficient 3/2 in the right side represents a coefficient whenrelative conversion is used as d-q coordinate conversion. In the case ofabsolute conversion, the coefficient becomes 1. Since the right side ofthe equation (8) is established in any coordinate, a relation on a dc-qcaxis can be represented by an equation (9). $\begin{matrix}{{I_{0}V_{0}} = {\frac{3}{2}( {{V_{d\quad c}I_{d\quad c}} + {V_{q\quad c}I_{q\quad c}}} )}} & \lbrack {{Equation}\quad 9} \rbrack\end{matrix}$

[0103] Assuming that the inverter is ideal, Vdc and Vqc can be replacedby Vdc* and Vqc*, and voltage commands can be used instead. Iqc obtainedfrom the equation (9) is represented by an equation (10).$\begin{matrix}{I_{q\quad c} = \frac{{\frac{2}{3}I_{0}V_{0}} - {V_{d\quad c}^{*}I_{d\quad c}}}{V_{q\quad c}^{*}}} & \lbrack {{Equation}\quad 10} \rbrack\end{matrix}$

[0104] In the equation (10), since Idc cannot be detected, if a commandvalue Id* is used instead, a relation is represented by an equation(11). $\begin{matrix}{I_{q\quad c} = \frac{{\frac{2}{3}I_{0}V_{0}} - {V_{d\quad c}^{*}I_{d}^{*}}}{V_{q\quad c}^{*}}} & \lbrack {{Equation}\quad 11} \rbrack\end{matrix}$

[0105] When the Idc is replaced by Id*, an estimation error may beslightly increased. However, since a q axis (qc axis) is dominant in anoutput of the synchronous motor, no large errors are generated. The Iqcestimator estimates Iqc by using calculation of the equation (11). A DCvoltage V0 may be directly detected by using a sensor. However, iffluctuation in a DC voltage is small, a set value (command value) of theDC current can be used.

[0106] In addition, since the equation (8) is a relational equation whenconversion efficiency of the inverter is assumed to be 1, the estimatedvalue includes an error due to the assumption. Thus, to increaseaccuracy of estimation, Iqc may be estimated by considering theconversion efficiency of the inverter.

[0107] According to the third embodiment, without using any currentsensors of the synchronous motor, it is possible to achieve asynchronous motor driving system of a vector control sensorless system,which is simpler in configuration, and higher in performance.

[0108] Fourth Embodiment

[0109]FIG. 8 is a block diagram showing an internal configuration of anaxial error calculator 14D in a synchronous motor driving systemaccording to a fourth embodiment of the present invention. In the fourthembodiment, instead of the axial error calculator 14 in the first orthird embodiment, the axial error calculator 14D including a functiongenerator 21 for generating a function shown in FIG. 9 is used.

[0110] Axial error calculation of the present invention can be achievedby removing the R term of the equation (4) as described above. In thismethod of calculation, when a rotational speed was extremely low, i.e.,1 to 2% or lower, there was a possibility of an error generated in aresult of axial error calculation. Thus, the equation of calculation iscorrected in order to increase accuracy of axial error calculation in avery low speed zone. By modifying the equation (4), a relation isrepresented by an equation (12). $\begin{matrix}{{\Delta\theta} = {\tan^{- 1}\frac{{L_{q}( {I_{q\quad c} - I_{q}^{*}} )} - \frac{R( {I_{d\quad c} - I_{d}^{*}} )}{\omega_{1}}}{K_{e} - ( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} ) - \frac{R( {I_{q\quad c} - I_{q}^{*}} )}{\omega_{1}}}}} & \lbrack {{Equation}\quad 12} \rbrack\end{matrix}$

[0111] In the equation (12), it can be understood that as ω1 is smaller,an effect of an R term is larger.

[0112] However, direct use of the equation (12) may cause a considerableincrease in a calculation error when ω1 is very small. In a worst case,division by zero may even occur. Thus, the equation (12) is modified asfollows: $\begin{matrix}{{\Delta \quad \theta} = {\tan^{- 1}\frac{{L_{q}( {I_{qc} - I_{q}^{*}} )} - {{K_{rx}( \omega_{1} )}( {I_{d\quad c} - I_{d}^{*}} )}}{K_{e} - ( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} )}}} & \lbrack {{Equation}\quad 13} \rbrack\end{matrix}$

[0113] Since a term of Ke is dominant in denominator term of theequation (12), correction is made only for a numerator by a functionKrx.

[0114]FIG. 9 illustrates an operation of a function generator in thesynchronous motor driving system of the fourth embodiment of the presentinvention. A function Krx is set, for example, similar to that shown inFIG. 9. In a region where ω1* is equal to/higher than ω1L, Krx=0 is set,and a term for R is removed. Only when ω1* is equal to/lower than ω1L,axial error calculation is corrected to increase accuracy of axial errorestimation. However, a function Krx at time of ω1*=0 is limited to afinite value (Krx0), and problems of calculation such as division byzero are prevented. As a result, even in the very low speed zone, it ispossible to increase the accuracy of axial error calculation, andachieve a synchronous motor driving system of a vector controlsensorless system in a wider range.

[0115] Fifth Embodiment

[0116]FIG. 10 is a block diagram showing an internal configuration of acontrol unit 2E in an synchronous motor driving system according to afifth embodiment of the present invention. In the fifth embodiment,instead of the Iq* generator 11 for supplying a current command Iq* of aq axis component (torque component) of the synchronous motor, an Iq*generator 11E is provided. The Iq* generator 11E of the fifth embodimentcalculates a q axis current command Iq* based on a detected currentvalue Iqc.

[0117] In the case of vector control, it is always necessary to controla voltage applied to the synchronous motor and a current of thesynchronous motor to a relation represented by the equation (2). Id* hasno direct relation to a load on the synchronous motor, and thus can beset to an optional value. On the other hand, Iq* must be properlychanged according to load torque, and a rotational speed.

[0118] In a stationary state, a relation of Iq*=Iqc must always be set.Otherwise, an axial error Δθ is left, making it impossible to establishvector control. According to the fifth embodiment of the presentinvention, by an extremely simple system configuration, Iq* can bematched with Iqc. The Iq* generator 11E calculates an equation (14).Here, a code Tr denotes a time constant; and s Laplacian operator.$\begin{matrix}{I_{q}^{*} = {\frac{1}{1 + {T_{r} \cdot s}} \cdot I_{qc}}} & \lbrack {{Equation}\quad 14} \rbrack\end{matrix}$

[0119] The equation (14) represents a first order lag element and, inprinciple, a stationary value of Iqc is set as Iq*. Accordingly, Iqc=Iq*is established at the end, establishing vector control.

[0120] A current command in normal control is supplied before a realdetected current value, and the detected value is matched with thecurrent command. However, in the system configuration of FIG. 10,different from the normal case, a command is later matched with anecessary current value, i.e., an actually flowing current value, andthereby balance is kept between a voltage and a current.

[0121] By using the control unit of FIG. 10, there are two places wherecontrol constants are adjusted, i.e., a magnetic pole axis estimationgain 17, and the time constant of the equation (14). The systemconfiguration is thus simplified greatly, realizing vector control.Moreover, since the axial error calculator 14 enables high-performancecontrol to be achieved in range of substantially all speeds, there isproblem of system switching for each speed zone, and it is possible toachieve a synchronous motor driving system of a vector controlsensorless system having the number of places to be adjusted set to aminimum.

[0122]FIG. 11 is a block diagram showing an internal configuration of acontrol unit 2F when the synchronous motor driving system of the fifthembodiment of the present invention is applied to the third embodimentof FIG. 7. In FIG. 11, an Iq* generator 11E fetches an estimateddetected current value not from the dq coordinate converter 9 of FIG. 10but from the Iqc estimator 20 of FIG. 7. Also in this case, a q axiscurrent command Iq* is calculated based on a detected current value Iqc.Thus, by a simpler system configuration, it is possible to achieve asynchronous motor driving system of a vector control sensorless system.

[0123] Sixth Embodiment

[0124]FIG. 12 is a block diagram showing an internal configuration of acontrol unit 2G in a synchronous motor driving system according to asixth embodiment of the present invention. The control unit 2G of thesixth embodiment includes an Id current control unit 22 for controllinga d axis current, an Iq current control unit 23 for controlling a q axiscurrent, a code inverter 24 for inverting a code of Δω1, a conversiongain 25 for converting Δω1 into speed deviation with P set as a pole ofthe synchronous motor, and a speed control unit 26 for setting speeddeviation to zero.

[0125] As described above with reference to the fifth embodiment, in thevector control, making of a current command Iq* is extremely important.In the fifth embodiment, since a current command Iq* is obtained from areal detected current value (estimated value), it is very easy. However,high-speed response to speed or load fluctuation is difficult.

[0126] On the other hand, by using the control unit 2G shown in FIG. 12,it is possible to achieve a synchronous motor system of a vector controlsensorless system capable of making high-speed response.

[0127] An output Δω1 of a magnetic pole axis estimation gain 17 is acorrected speed amount for reducing an axial error to zero. In otherwords, this output is an amount corresponding to deviation between areal rotational speed command ωr* and a real speed ωr. Thus, bysupplying Iq* so as to reduce the output Δω1 to zero, response of thespeed control of the synchronous motor can be improved.

[0128] The output Δωl has its polarity inverted by the code inverter 24,multiplied by 2/P by the conversion gain 25, and speed deviation Δωr(=ωr*−ωr) is obtained. The speed control unit 26 includes aproportional/integration compensation element, and the like, andcalculates a torque current command Iq* based on the speed deviationΔ107 r.

[0129] Moreover, to improve a response characteristic of the synchronousmotor, current control units 22 and 23 are added to dc, and qc axes, anda current is controlled at a high speed.

[0130] As a result, it is possible to achieve a synchronous motordriving system capable of making high-speed response to speedfluctuation, external torque disturbances, and the like. In addition,since an axial error calculator can be applied in a range ofsubstantially all speeds, it is possible to achieve a synchronous motordriving system of a vector control sensorless system having controlperformance considerably improved compared with the conventional vectorcontrol sensorless system.

[0131] Seventh Embodiment

[0132]FIG. 13 is a block diagram showing an internal configuration of acontrol unit 2H in a synchronous motor driving system according to aseventh embodiment of the present invention. In FIG. 13, a magnetic poleaxis estimation gain 17H includes a proportion/integration compensationelement. The control unit 2H of FIG. 13 is substantially similar insystem configuration to the control unit 2G of the sixth embodiment, butdifferent in a method of making ω1 in the control unit 2H.

[0133] In the foregoing first to sixth embodiments, ω1* was directlycalculated from the rotational speed command ωr*, Δω1 was added, and thedriving frequency was corrected.

[0134] On the other hand, the seventh embodiment has a feature that anoutput of a magnetic pole axis estimation gain is set to be ω1, used forcontrol calculation.

[0135] A speed control unit 26 calculates Iq* based on deviation Δωbetween a rotational speed command ωr* and a real speed (estimatedvalue) ωr. The Iq* is compared with a real current value Iqc, and acurrent is controlled by an Iq current control unit 23 such that bothcan be matched with each other. When a torque current is actuallygenerated in the synchronous motor, and a rotational speed of thesynchronous motor is changed to generate an axial error, an axial errorcalculator 14 detects this axial error. The magnetic pole axisestimation gain 17H receives the axial error, corrects the ω1 andoutputs it.

[0136] With the control system constructed in such a manner, it ispossible to perform acceleration/deceleration of the synchronous motorby maximum torque. A current to be supplied to the synchronous motor islimited by synchronous motor rating or an inverter capacity. Byperforming acceleration/deceleration by a maximum condition in thisrange, it is possible to control the synchronous motor at a highestspeed.

[0137] Accordingly, a limiter of a maximum current is provided in theIq* and, in a maximum flowing state of a torque current, the synchronousmotor is accelerated. In this case, a speed ω1 must be a speed resultedfrom application of torque and, different from the case shown in FIG.12, a speed cannot be provided from a rotational speed command in a feedforward manner. Thus, when the synchronous motor isaccelerated/decelerated by maximum torque, the system configuration ofFIG. 13 is necessary.

[0138] By employing the axial error calculator 14 of the seventhembodiment, it is possible to achieve a high-speedacceleration/deceleration characteristic in a wide range.

[0139] The control units 2G and 2H of FIGS. 12 and 13 can be applied tothe system of a configuration shown FIG. 5 or 7. If applied to thesystem configuration of FIG. 5, the control unit only needs to bedirectly replaced. If applied to the system configuration of FIG. 7, thefilter 19 and the Iqc estimator 20 need to be provided in the controlunit. However, since Idc cannot be detected in the system configurationof FIG. 7, no Id current control units are installed.

[0140] According to the seventh embodiment, it is possible to achieve asynchronous motor driving system having control performance considerablyimproved compared with a high-performance sensorless system havingfurther improved high-speed responsiveness.

[0141] Eighth Embodiment

[0142]FIG. 14 is a block diagram showing an internal configuration of acontrol unit 2J in a synchronous motor driving system according to aneighth embodiment of the present invention. In the eighth embodiment,instead of the Id* generator 10 in the first to seventh embodiments, anId* generator 10J is used. The Id * generator 10J of FIG. 14 decides avalue of Id* based on Iq*. That is, the eighth embodiment issubstantially similar to the control unit of the fifth embodiment shownin FIG. 10, but has a feature in a method of making a current commandId*.

[0143] Among permanent magnet synchronous motors, there is a type, whichgenerates synchronous motor torque by combining torque generated by apermanent magnet with reluctance torque generated by saliency (reversesaliency) of the synchronous motor. In the case of the synchronous motorof this type, a maximum torque point of the synchronous motor is locatedin a region where Id is controlled to be a minus value, and control ofId=0 is not advantageous for efficiency. Thus, to drive the synchronousmotor by maximum efficiency, the synchronous motor is preferably drivenalways by maximum torque. Especially, in an industrial/home electricappliance field, energy conservation has been demanded, and maximizationof efficiency is an important task.

[0144] Conditions for obtaining maximum torque are described in, forexample in a document 3: pp. 662-667, “Comparison of ControlCharacteristics of Permanent Magnet Synchronous Motors with SeveralRotor Configurations”, JIEE papers D, Vol. 114-6, 1994, or the like.According to an equation (6) of the document 3, a relation isrepresented by an equation (15). When Iq is set, Id for obtainingmaximum torque is decided. However, Φm denotes a magnetic flux of apermanent magnet, and Ld≠Lq is established. $\begin{matrix}{I_{d} = {\frac{\Phi_{m}}{2( {L_{q} - L_{d}} )} - \sqrt{\frac{\Phi_{m}^{2}}{4( {L_{q} - L_{d}} )^{2}} + I_{q}^{2}}}} & \lbrack {{Equation}\quad 15} \rbrack\end{matrix}$

[0145] In the eighth embodiment, the Id* generator 10J calculates theequation (15) by using Id*. As a result, it is always possible to drivethe synchronous motor by maximum torque (maximum efficiency). For thecalculation of the equation (15), Iqc may be used instead of Id*.However, since fluctuation is large in the Iqc during transition, theentire control system may become unstable.

[0146] Efficiency maximization can contribute to energy conservation ofthe apparatus if functioning in s stationary state, and thus use of Iq*as an output of the Iq* generator causes no problems.

[0147] Moreover, application of the Id* generator 10J of the eighthembodiment to the other first to seventh embodiments causes no problems.

[0148] Thus, by using the eighth embodiment, it is possible to provide asynchronous motor system of a vector control sensorless system, which iscapable of operating the synchronous motor by maximum efficiency.

[0149] According to the present invention, it is possible to provide thesynchronous motor driving system capable of achieving the vector controlsensorless system to cover the wide range from the low to high speedzone without any reductions in efficiency or increases in noise.Moreover, irrespective of presence of saliency of the synchronous motorto be controlled, it is possible to achieve the synchronous motordriving system of the high-performance, and highly accurate vectorcontrol sensorless system.

[0150] It should be further understood by those skilled in the art thatthe foregoing description has been made on embodiments of the inventionand that various changes and modifications may be made in the inventionwithout departing from the spirit of the invention and the scope of theappended claims.

What is claimed is:
 1. A synchronous motor driving system whichcomprises a synchronous motor, an inverter for driving the synchronousmotor, a rotational speed command generator for supplying a rotationalspeed command to the synchronous motor, and a control unit forcalculating a voltage applied to the synchronous motor, said synchronousmotor driving system comprising axial error calculation means forestimating an axial error Δθ between a d-q axis and a dc-qc axis byusing Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of all rotationalspeeds except zero of the rotational speed command of the synchronousmotor wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, Idc is a detected currentvalue on an assumed dc axis on control, and Iqc is a detected currentvalue on an assumed qc axis orthogonal to the assumed dc axis; and meansfor adjusting the dc-qc axis to the d-q axis based on the calculatedvalue of the axial error Δθ.
 2. A synchronous motor driving system whichcomprises a synchronous motor, an inverter for driving the synchronousmotor, a rotational speed command generator for supplying a rotationalspeed command to the synchronous motor, and a control unit forcalculating a voltage applied to the synchronous motor, said synchronousmotor driving system comprising axial error calculation means forestimating an axial error Δθ as a function of an inductance L and agenerated power constant Ke among the resistance R, the inductance L anda generated power constant Ke as synchronous motor constants of thesynchronous motor in a range of all rotational speeds except zero of arotational speed command of the synchronous motor; and means foradjusting the dc-qc axis to the d-q axis based on the calculated valueof the axial error Δθ, wherein Ld is an inductance on a magnetic poleaxis d, Lq is an inductance on a q axis orthogonal to the magnetic poleaxis d, Ke is a generated power constant of the motor, Id* is a currentcommand of the d axis, Iq* is a current command on the q axis, Idc is adetected current value on an assumed dc axis on control, and Iqc is adetected current value on an assumed qc axis orthogonal to the assumeddc axis.
 3. The synchronous motor driving system according to claim 2,wherein the axial error calculation means is means for calculating anaxial error Δθ in accordance with an equation (1) by using currentcommands Id* and Iq* on the d-q axis, and detected current values Idcand Iqc on the dc-qc axis, wherein Ld is an inductance on a magneticpole axis d, Lq is an inductance on a q axis orthogonal to the magneticpole axis d, Ke is a generated power constant of the motor, Id* is acurrent command of the d axis, Iq* is a current command on the q axis,Idc is a detected current value on an assumed dc axis on control, andIqc is a detected current value on an assumed qc axis orthogonal to theassumed dc axis: $\begin{matrix}{{\Delta \quad \theta} = {\tan^{- 1}\frac{L_{q}( {I_{qc} - I_{q}^{*}} )}{K_{e} - ( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} )}}} & \lbrack {{Equation}\quad 1} \rbrack\end{matrix}$


4. The synchronous motor driving system according to claim 3, wherein acurrent command Id* is used, instead of the detected current value Idcon the dc axis.
 5. The synchronous motor driving system according toclaim 1, further comprising means for detecting a DC current on a powersource side of the inverter, and synchronous motor current estimatingmeans for estimating an AC current of the synchronous motor based on thedetected DC current and a driving pulse signal for driving the inverter,the axial error Δθ being calculated using the estimated current as adetected current value.
 6. The synchronous motor driving systemaccording to claim 2, further comprising means for detecting a DCcurrent on a power source side of the inverter, and synchronous motorcurrent estimating means for estimating an AC current of the synchronousmotor based on the detected DC current and a driving pulse signal fordriving the inverter, the axial error Δθ being calculated using theestimated current as a detected current value.
 7. The synchronous motordriving system according to claim 1, further comprising means fordetecting a DC current on a power source side of the inverter, and Iqcestimating means for estimating a current value on the qc axis of thesynchronous motor based on the detected DC current, and a detected valueor a set value of a DC voltage of the inverter, the axial error Δθ beingcalculated using the estimated current as a detected current value. 8.The synchronous motor driving system according to claim 2, furthercomprising means for detecting a DC current on a power source side ofthe inverter, and Iqc estimating means for estimating a current value onthe qc axis of the synchronous motor based on the detected DC current,and a detected value or a set value of a DC voltage of the inverter, theaxial error Δθ being calculated using the estimated current as adetected current value.
 9. The synchronous motor driving systemaccording to claim 1, further comprising a correction term for makingcorrection in accordance with a rotational speed command of thesynchronous motor in the calculation of the axial error Δθ, saidcorrection term being a function of weight which increases as therotational speed command approaches zero.
 10. The synchronous motordriving system according to claim 2, further comprising a correctionterm for making correction in accordance with a rotational speed commandof the synchronous motor in the calculation of the axial error Δθ, saidcorrection term being a function of weight which increases as therotational speed command approaches zero.
 11. The synchronous motordriving system according to claim 1, wherein the current command Iq* onthe q axis is made based on the detected current value or the estimatedvalue on the qc axis.
 12. The synchronous motor driving system accordingto claim 2, wherein the current command Iq* on the q axis is made basedon the detected current value or the estimated value on the qc axis. 13.The synchronous motor driving system according to claim 1, furthercomprising means for estimating a speed deviation between the rotationalspeed command and a real rotational speed based on the calculated valueof the axial error Δθ, the q axis current command Iq* of the synchronousmotor being made based on the estimated value of the speed deviation.14. The synchronous motor driving system according to claim 2, furthercomprising means for estimating a speed deviation between the rotationalspeed command and a real rotational speed based on the calculated valueof the axial error Δθ, the q axis current command Iq* of the synchronousmotor being made based on the estimated value of the speed deviation.15. The synchronous motor driving system according to claim 1, furthercomprising means for estimating a rotational speed of the synchronousmotor based on the calculated value of the axial error Δθ, the q axiscurrent command Iq* of the synchronous motor being made based on adeviation between the estimated value and the rotational speed command.16. The synchronous motor driving system according to claim 2, furthercomprising means for estimating a rotational speed of the synchronousmotor based on the calculated value of the axial error Δθ, the q axiscurrent command Iq* of the synchronous motor being made based on adeviation between the estimated value and the rotational speed command.17. The synchronous motor driving system according to claim 11, whereinthe current command Id* of the d axis is made based on the currentcommand Iq* of the q axis.
 18. The synchronous motor driving systemaccording to claim 12, wherein the current command Id* of the d axis ismade based on the current command Iq* of the q axis.
 19. The synchronousmotor driving system according to claim 13, wherein the current commandId* of the d axis is made based on the current command Iq* of the qaxis.
 20. The synchronous motor driving system according to claim 14,wherein the current command Id* of the d axis is made based on thecurrent command Iq* of the q axis.
 21. The synchronous motor drivingsystem according to claim 15, wherein the current command Id* of the daxis is made based on the current command Iq* of the q axis.
 22. Thesynchronous motor driving system according to claim 16, wherein thecurrent command Id* of the d axis is made based on the current commandIq* of the q axis.
 23. The synchronous motor driving system according toclaim 1, wherein said synchronous motor is of a non-salient type. 24.The synchronous motor driving system according to claim 2, wherein saidsynchronous motor is of a non-salient type.